Analog to digital converter calibration via synchronous demodulation

ABSTRACT

A technique for dynamically calibrating a successive approximation charge to digital converter by toggling at least some portion of the converter between two predetermined states, with the design goal of balancing the voltage and/or charge that is output in the two states. The two states are chosen such that they are expected to generate the same output voltage when the converter is in “normal” operation mode, e.g., within a fraction of the Least Significant Bit (LSB) resolution of the converter. If there is an imbalance, switching between the two calibration states invariably generates a square wave signal that toggles between two distinct values. A synchronous demodulator having a bandwidth centered at the toggle frequency can then be used to accurately detect an amount of error, which is then feedback to generate correction signals. If there are undesirable static offsets introduced by the synchronous demodulator or by the signal and/or charge levels output by the two differential halves of the converter, a properly timed latch can be used to further stabilize the error signal.

RELATED APPLICATION(S)

This application continuation of U.S. application Ser. No. 10/870,330,filed Jun. 17, 2004. The entire teachings of the above application(s)are incorporated herein by reference.

BACKGROUND OF THE INVENTION

The present invention relates to calibrating Analog-to-DigitalConverters (ADCs) or Digital-to-Analog Converters (DACs), especiallythose which use Charge Coupled Device (CCD) pipeline structures andsuccessive approximation techniques.

Many modern electronic systems require conversion of signals from analogto digital or from digital to analog form. Circuits for performing thesefunctions are now required in numerous common consumer devices such asdigital cameras, cellular telephones, wireless data network equipment,audio devices such as MP3 players, and video equipment such as DigitalVideo Disk (DVD) players, High Definition Digital Television (HDTV)equipment, and numerous other products.

U.S. Pat. No. 4,375,059 issued to Schlig is an early example of a ChargeCoupled Device (CCD) based converter. In that design, a number of chargestorage stages are arranged as a serial pipeline register so that aninput source charges pass from stage to stage down the pipeline. Areference charge generator and a charge splitter at each stage generatereference signals. A first of the reference signals is compared to asource charge that is temporarily stored at the stage. The comparisongenerates a binary one if the source charge is greater than or equal tothe first reference charge, or a binary zero if this source charge isless than the first reference charge. If a binary one is generated, onlythe stored contents of the stage need pass through to the nextsuccessive stage. However, if a binary zero is generated, the storedcontents of the stage are passed to a next successive stage, togetherwith a second reference charge, in such a way that the stored chargesare combined. Auxiliary buffer registers are provided to temporarilystore the output bits of the comparators. This allows forming a digitalword for each source charge packet as the packet and its associatedcharge components travel down the pipeline.

A further refinement in charge to digital converter design is found inU.S. Pat. No. 5,579,007 issued to Paul. In that arrangement, thepipeline produces a serial stream of both positive and negative signalcharges corresponding to a differential signal. The differential signalstructure provides improved sensitivity in the charge to voltagetranslation process, and thus increased dynamic range. The structurealso exhibits reduce sensitivity to mismatches, by suppression of commonmode noise signals in the charge domain.

In order to provide a high precision converter, the differential typesuccessive approximation pipeline must often be trimmed or calibrated.The precision of the calibration apparatus must therefore beconsiderably better than the converter itself, making its design quitechallenging.

Existing converter calibration techniques typically set the converter toa static state and then adjust one or more parameters of the pipeline toprovide for Direct Current (DC) balance. These techniques usuallyrequire precise, low noise, low DC-offset amplifiers and/or comparators.Unfortunately, thermal noise, low frequency (l/f) noise, and DC voltageoffsets produced by these devices often limit how accurately theconverter can be calibrated.

SUMMARY OF THE INVENTION

The present invention is a technique for dynamically calibrating asuccessive approximation charge to digital converter by toggling atleast some portion of the converter between two predetermined states,with the design goal of balancing the voltage and/or charge that isoutput in the two states. When the converter is out of balance, thevoltages differ, producing an error signal.

In other words, the two states are chosen such that they are normallyexpected to generate the same output voltage, within a fraction of theaccuracy of the Least Significant Bit (LSB) of the converter. If thereis an imbalance, switching between the two states invariably generates asquare wave signal that toggles between two distinct values. The errorsignal itself changes state at the toggle rate. A synchronousdemodulator having a bandwidth centered at the toggle frequency can thenbe used to accurately detect the amount of error, even in the presenceof significant Direct Current (DC) voltage offsets and low frequency(l/f) noise. The synchronous demodulator can be designed to be verynarrow band, rejecting both low and high frequency signals as well asnoise and DC offsets.

In a preferred embodiment, the synchronous demodulator is implementedwith a mixer and a low pass filter. The mixer receives the error signaland a signal corresponding to the toggle rate. The low pass filter maybe implemented with an integrator. In the case of small error signalamplitudes, this approach can obtain improved performance by increasingthe integrator time constant.

If there are undesirable static offsets introduced by the synchronousdemodulator itself or in the signal and/or charge levels output by thetwo differential halves of converter, a sawtooth waveform will result.This sawtooth ends up being superimposed on the normally linear rampproduced by the integrator. In further embodiments, therefore, aproperly timed latch is coupled to the integrator output to remove theeffect of the offset on the ramp. The latch ensures that the errorsignal is only sampled after a complete sawtooth up-down cycle time.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, features and advantages of theinvention will be apparent from the following more particulardescription of preferred embodiments of the invention, as illustrated inthe accompanying drawings in which like reference characters refer tothe same parts throughout the different views. The drawings are notnecessarily to scale, emphasis instead being placed upon illustratingthe principles of the invention.

FIG. 1 is a high level block diagram of a charge-to-digital converterthat uses the techniques of the present invention.

FIG. 2 is a more detailed view of the converter pipeline.

FIG. 3 illustrates a set of switch control signals applied to thecomponents of the converter pipeline to implement two states, eachhaving nominally the same output value.

FIG. 4A illustrates a differential output signal for the two states, fortwo different operating conditions.

FIG. 4B illustrates the synchronous demodulator output for the twoconditions.

FIG. 5 is a more detailed view of an alternate embodiment of thesynchronous demodulator combined with a chopper-stabilized amplifier.

FIG. 6 is a signal diagram showing the integrator output before andafter being latched.

FIG. 7 illustrates a ring counter implementation for the switch controlsignal generator.

DETAILED DESCRIPTION OF THE INVENTION

A description of preferred embodiments of the invention follows.

FIG. 1 is a high level illustration of a converter system 100 thatoperates according to principals of the present invention. The system 10consists of an input switch 110, a converter core 120, state signalgenerator 130, a synchronous demodulator 140, and a processor 150. Ingeneral, the present invention is a technique for dynamicallycalibrating the converter core 120, which may be a successiveapproximation charge to digital converter, or some other voltageconverter, by toggling at least some portion of the converter betweentwo predetermined, nominally balanced, states (state A and state B).

The input switch 110 provides a converter input signal (IN) to aconverter core 120. The converter input, IN, may be selected from eithera system input voltage (INPUT), such as when the system 10 is running inits normal operating mode, or as a reference voltage (V_(REF)), such aswhen the system 10 is running in a calibration mode. The particularoperating or calibration mode is selected by a switch control inputsignal, CAL.

In one preferred embodiment, the converter core 120 may have twophysical signal paths 125-1 and 125-2 that represent the two states,state A and state B. In other preferred embodiments, the converter core120 may have a single signal path that operates in two different modesto provide the two different states.

When in a calibration mode, the system 10 acts as a feedback loop thathas a goal of balancing the output between the two predetermined states.When the converter core 120 is out of balance, the voltages and/orcharges in the two states differ, producing an error signal 145.

The two states are chosen such that they are normally expected togenerate the same output charge, OUT, at the output of the convertercore 120. However, if the converter is not perfectly balanced, switchingbetween the two states generates a square wave signal OUT that actuallytoggles between two values, as either of the two paths are alternatelyselected by output switch 128.

The synchronous demodulator 140 has a bandwidth centered at the stateA/B toggle frequency. Thus, the synchronous demodulator 140 can be usedto accurately detect the amount of error, even in the presence ofsignificant Direct Current (DC) voltage and low frequency (l/f) noise.

In preferred embodiments, the synchronous demodulator 140 can be a mixerand an integrator. The mixer 141 is typically a multipler orcross-coupled switches. The integrator 142 controls the bandwidth of thedemodulator 140. It can be designed to be very narrow band, to rejectboth low and high frequency signals, as well as to reject noise and DCoffsets.

The integrated value of the error signal can in turn be further latched143, prior to the error signal 145 being fed to the processor 150. Theprocessor 150 then uses the error signal to apply control signals 160that adjust the operation of the two signal paths in the converter core.

FIG. 2 is a more detailed diagram of one embodiment of the invention asapplied to a so-called Charge Domain Converter (QDC) system 200operating as an Analog to Digital Converter (ADC). This particular QDC200 is a successive approximation type converter that uses a number ofcharge storage stages arranged as a serial pipeline register, so that aninput source charge passes from stage to stage down the pipeline. Areference charge generator and a charge splitter at each stage generatereference signals which are optionally added to the charge as it travelsdown the pipeline. In the illustrated embodiment, there are actually twopipelines 230-1, 230-2 that produce a serial stream of both positive andnegative signal charges corresponding to a differential signal at thepipeline outputs. The complimentary outputs are then fed to adifferential amplifier 235. This converter core is implemented along thelines of the QDC described in the aforementioned U.S. Pat. No. 5,579,007issued to Paul.

More particularly, an input voltage to be converted is presented as acomplimentary pair of voltages, Vinp and Vinm, representing a positive(plus) and negative (minus) version of the input signal to be converted.Switches 220-1, 220-2, one for each of the plus and minus paths throughthe converter core, provide a selected input signal to a respectivesampler, 222-1, 222-2. The samplers 222 each convert a respective inputvoltage to a charge. In normal operation mode of the QDC 200, theseinput signals are selected by the switches 220. In the calibration mode,however, the same common mode voltage, Vcm, is fed through the samplers222 to each of the converter pipelines 230. In the preferred embodiment,Vcm, is equal to one-half the full scale input value. This is an inputcondition that results in the same nominal output value being providedby each of the plus and minus paths.

The charges output by the samplers 222-1, 222-2 are fed to the inputstage of the respective charge pipeline 230-1, 230-2, which areimplemented as Charge Coupled Device (CCD) type analog shift registers.Each of the plus and minus paths through the converter core 210 have arespective digital-to-analog (DAC) ladder DACp (230-1) or DACm (230-2).

Each ladder consists of reference charge generator 225-1, 225-2 and aseries of adjustable charge splitters 226 (not individually numbered forthe sake of clarity). There is an adjustable charge splitter 226associated with each stage of each of the pipelines 230. The chargesplitters 226 are arranged in series to couple (or, to not couple,depending upon the setting of a corresponding switch 227) a fractionalamount of charge to the respective stage of the pipeline 230. Eachsuccessive splitter 226 provides one-half of the reference charge itreceives to the next splitter in the chain. Thus, the string ofsplitters provide an amount of charge equal to ½, ¼, ⅛, 1/16, . . . ,1/2 ^(i) (where i is the number of stages in the pipeline) of thereference charge.

In normal operation of the converter core, a set of fast comparators229, one for each stage of the pipeline (not shown in detail), providean analog charge-to-digital conversion result.

However, of more interest to the present invention is operation in thecalibration mode. In that mode, the switches 227 are controlled insteadby a set of switch control signals 232 provided by a digital shiftregister 230. In the calibration mode, the switches 227 are configuredso that the converter core may be operated in one of two states, state Aor state B, that nominally each provide the same output charge. Thesystem is operated in the calibration mode such that it changes betweenstate A and state B at a calibration or “A/B” toggle rate. The A/Btoggle rate can be any convenient frequency at which the components ofthe system operate properly.

Please note also that in the calibration mode an extra bit of thepipeline, beyond the Least Significant Bit (LSB) used in normaloperation, is enabled for use. The extra stage beyond the nominal LSB,as will be understood shortly, provides the capability for producing twonominally equal output states, within the range of the LSB resolution ofthe converter, but by actually using two different input states.

FIG. 3 illustrates the state of the switches 227 more particularly. Asalluded to previously, the switches 227 are used to control theconverter core so that it is toggled between two different states, stateA and state B, that are expected to provide the same output value. InFIG. 3 a convention is adopted such that a binary 1 represents a switchcontrol signal that places its corresponding switch 227 in the closedstate, with a binary 0 representing a switch control signal that opensits corresponding switch.

In a first state A, illustrated in the top two lines of FIG. 3, the plusladder or DACp is fed control signals 1 0 0 . . . 0 [0], with logic 1being fed to control the switch for the first splitter 226 in thepipeline 230-1 (the one receiving ½ the reference charge), so that onlythat stage is permitted to feed charge to the pipeline 230-1. Pleasenote a convention here of the bracketed value [0] indicating the logicstate associated with the added bit which is the stage i+1^(th) bit,beyond the LSB stage i. Also in state A, the minus ladder or DACm is fedcontrol signals 0 1 1 . . . 1 [1], to allow all stages in pipeline 230-2except the first to receive charge. Thus, in this state A, thedifferential amplifier 235 provides an output corresponding to thedifference between these two input settings 1 0 0 . . . 0 [0] and 0 1 1. . . 1 [1].

The additional bracketed bits provide two states that are capable ofproducing the “same” output, to an accuracy of a fraction of theconverter's LSB resolution.

State B, illustrated in the bottom two lines of FIG. 3, represents adifferent state of the converter core that nominally provides the sameoutput. In state B, the plus ladder DACp is fed control signals 0 1 1 .. . 1 [1], and the minus ladder DACm is fed control signals 1 0 0 . . .0 [0]. Thus, in this state B, the differential amplifier 235 provides anoutput corresponding to the difference between these two input settings0 1 1 . . . 1 [1] and 1 0 0 . . . 0 [0].

In the circuit of FIG. 3, these switch control signals are shown beinggenerated by feeding a square wave at the A/B toggle clock frequency.These signals might be generated by a clock divider associated with eachcontrol line running at the A/B rate. However, a particular preferredembodiment of a switch control signal generator that uses a ring counteris useful when A/B is related to the clock frequency. Thisimplementation for the switch control signal generator will be describedbelow in connection with FIG. 7.

While the outputs in states A and B should nominally be the same, andthus the output of the differential amplifier 235 should be a constantvalue, in fact the output is a square wave, as illustrated, due todifferences in the calibration of DACp and DACm.

As previously mentioned, mixer 240 and integrator 241 operate as asynchronous demodulator (driven by the A/B state signal fed to mixer240) to detect the error signal and to drive the integrator 241 outputto one voltage rail or the other.

FIG. 4A illustrates a typical output of the integrator 241 for the plusladder DACp, for two different operating conditions. A first condition,illustrated by the solid line square wave signal 401, alternates betweentwo values at the A/B toggle rate. The difference in output betweenstate A and state B under these conditions is Δ1. The dotted line squarewave 402 illustrates the output under a different set of operatingconditions, where the difference in output is Δ2.

FIG. 4B illustrates the output of the integrator 241 for the twoconditions. In the first instance, illustrated by the solid line rampsignal 403, the integrator output ramps to a voltage rail at time t1with slope s1. However, in the case of dotted line 404, the integratoroutput ramps more slowly with slope s2, to time t2. The synchronousdemodulator 240 thus converges rapidly when the differential A/B outputis large. However, it also permits resolution of smaller differences inA/B state output, by simply allowing a longer integration time.

FIG. 5 illustrates a phenomenon of the integrator and comparator in moredetail, and the preference for including latch 242. In particular,consider a simple situation where there is no offset in the calibrationcircuitry, and the pipelines 230 provide a perfectly constant output inthe calibration mode. In this scenario, both complimentary outputs ofthe integrator, V+ and V−, will appear as an accumulation of thisconstant voltage, that is, as continuous ramp up and ramp down signals051 and 502, as indicated by the dotted lines with slope s1.

However, consider when the components of the system introduce an offset.There are several possible sources for the offset, either in thesynchronous demodulator itself, or more likely, differences introducedby the two paths 125-1, 125-2 through the converter core. As indicatedby the solid line signals 503 and 504, the simple ramps now have afurther sawtooth-like modulation impressed upon them, with the period ofthe sawtooth corresponding to the A/B toggle rate. Eventually, at timet4, the ramp will remain above the impressed sawtooth. The ramp willactually oscillate above and below sawtooth for a while after time t3,producing an unambiguous error signal. By adding the latch 243 timed tosample the sawtooth on the edge of the A/B clock periods, this effectcan be minimized. This is evident from the timing diagrams for thecomparator output (COMP OUT), and latch output (LATCH OUT), shows thelatch-stabilized error signal 145.

Processor 150 then receives the output of latch 243 and determines thevalues of adjustment signals applied to the adjustable splitters 230. Ifthe latch output is equal to a logic 1, then the processor 150 will setthe values of these signals to cause one or more of the splitters 230 toadjust a certain amount in one direction. If the latch output is equalto a logic value ∅, then the signals are set to values that adjust oneor more splitters 230 in the other direction. The processor 150 may useany convenient algorithm to converge to a splitter adjustment solution,including, but not limited to a binary search or a linear search.

Certain converter configurations advantageously make use of chopperstabilization for normal mode operation of the differential amplifier235. With this type of amplifier, DC offsets and low frequency l/f noiseare essentially removed, because the signal is shifted above DC. Inthese configurations, some of the circuitry used in a typicalchopper-stabilized amplifier can be used to implement portions of thesynchronous demodulator.

FIG. 6 is a schematic diagram illustrating one such possible embodiment.As before, the outputs of the DACm and DACp channels are provided by thebuffer amplifiers 232-1 and 232-2, respectively. The chopper stabilizedamplifier 600 consists of mixer 633, differential amplifier 635 andoutput mixer 640. To stabilize the converter output in normal operationmode, the first mixer 633 up-converts any DC signal received from theinput buffer amplifiers 232-1 and 232-2, up to some predeterminedcarrier. Differential amplifier 635 in turn operates to take adifference at a frequency higher than DC, avoiding the introduction offurther DC noise or low frequency l/f noise. The output mixer 240 thendown-converts the differential amplifier output back to DC.

It is therefore quite evident that a synchronous demodulator needed forthe calibration mode shares many components of the chopper stabilizedamplifier 600 used in the normal operation mode. In other words,amplifier 635 can serve to function as the high speed differentialamplifier 235 need for the calibration mode of FIG. 2, and mixer 640 canserve as the mixer 240. The first mixer 633 and second mixer 640 areeach fed the A/B toggle signal. The additional mixer 633 on the inputcan be set to merely pass through the signals output by the buffers232-1, 232-2. In the case of a multiplier implementation, it can be setto multiply by a value of 1.

Finally, FIG. 7 shows a more detailed view of a possible refinement forthe switch control signal generator. If the pipeline has 36 stages, forexample, 36 flip flops would be required to provide the control signalsif a simple shift register is used. Careful study of the sequence ofbits produced by the digital shift register 230 reveal that they areactually square waves of differing phases. For a converter with manypipeline stages, the control signals can be more efficiently generatedby a ring divider instead of many flip-flops, and prudent selection ofthe A/B toggle rate.

For example, if the A/B toggle rate is 1/16^(th) of the pipeline clockrate, then all phases of the required 8-bit on then 8-bit off waveformscan be generated with a single ring counter having only 8 flip flops,assuming that each flip flop provides true and complimentary outputs.The necessary control signals can then be selected from the proper phaseoutput of the ring counter, regardless of how many pipeline stages thereare.

While this invention has been particularly shown and described withreferences to preferred embodiments thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the scope of the inventionencompassed by the appended claims.

1. A digital converter apparatus comprising: a converter core having at least two predetermined calibration states in a calibration mode, each of the predetermined calibration states providing a respective predetermined output signal; a control circuit for switching the converter core between at least two of the predetermined calibration states; a sampler for providing a converter output signal over a sequence of multiple predetermined calibration states; a synchronous demodulator for demodulating the converter output signal wherein the control circuit further switches between the calibration states at a state toggle frequency, and a bandwidth of the synchronous demodulator is centered at the state toggle frequency.
 2. The apparatus of claim 1 additionally comprising: a correction circuit for using the demodulated converter output signal to provide a correction signal to the converter core.
 3. The apparatus claim 1 wherein the digital converter is a successive approximation converter.
 4. The apparatus of claim 1 wherein the converter core is a complimentary converter having two conversion signal paths therein, with a first conversion signal path operating as a plus signal path and a second path operation as a minus signal path.
 5. The apparatus of claim 4 wherein a first one of the predetermined calibration states is provided by applying a first set of calibration inputs to the plus signal path and a second set of calibration inputs to the minus signal path; and a second one of the predetermined calibration states is provided by applying the same second set of calibration inputs to the plus signal path and the same first set of calibration inputs to the minus signal path.
 6. The apparatus of claim 1 wherein the synchronous demodulator further comprises: an integrator for integrating the converter output signal to provide an error signal.
 7. The apparatus of claim 1 wherein each of predetermined calibration states provide nominally the same converter output signal.
 8. The apparatus of claim 1 additionally comprising: a chopper stabilizer connected to stabilize at least one component of the digital converter during an operating mode.
 9. The apparatus of claim 8 wherein the chopper stabilizer is additionally coupled to the synchronous demodulator.
 10. An apparatus for calibrating a digital converter comprising: a controller for toggling at least some portion of the converter between at least two predetermined calibration states in a calibration mode, with each of the two predetermined calibration states providing a predetermined output signal; an output circuit for providing a converter output signal over a sequence of multiple predetermined calibration states; a demodulator for synchronously demodulating the converter output signal; and an integrator for integrating the converter output signal to provide an error signal, wherein the error signal is generally a ramp type waveform in the absence of noise.
 11. The apparatus of claim 10 wherein offset voltages introduced by the converter superimpose a sawtooth type waveform on the ramp waveform.
 12. The apparatus of claim 11 additionally comprising: a latch for producing the error signal in synchronism with the change of calibration states of the converter.
 13. The apparatus of claim 10 wherein the converter is a complimentary converter having two conversion signal paths therein, with a first conversion signal path operating as a plus signal path and a second path operation as a minus signal path.
 14. The apparatus of claim 13 wherein a common reference input, Vcm, is fed to the plus signal path and the minus signal path.
 15. The apparatus of claim 10 wherein a number of stages used by the digital converter in the calibration mode is at least one stage greater than the number of stages of the converter used in an operation mode.
 16. An apparatus for calibrating a digital converter comprising: a controller for toggling at least some portion of the converter between at least two predetermined calibration states in a calibration mode, with each of the two predetermined calibration states providing a predetermined output signal; an output circuit for providing a converter output signal over a sequence of multiple predetermined calibration states; a demodulator for synchronously demodulating the converter output signal; wherein the converter is a complimentary type converter having a first conversion signal path operating as a plus signal path, and a second conversion signal path operating as a minus signal path; and wherein the two conversion signal paths are Charge Coupled Device (CCD) pipeline stages.
 17. The apparatus of claim 16 wherein calibration inputs are fed to a respective set of two or more adjustable charge splitters, with a charge splitter associated with a corresponding one of the CCD pipeline stages.
 18. The apparatus of claim 17 additionally comprising: an error circuit for generating an error signal from the synchronously demodulated converter output signal; and a splitter controller for providing adjustment signals to the adjustable splitters from the error signal. 